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19-1218; Rev 1; 6/97
MANUA ION KIT ALUAT EV BLE AVAILA
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Dual, High-Efficiency, PFM, Step-Up DC-DC Controller
____________________________Features
o Smallest Dual Step-Up Converter: 16-Pin QSOP o 90% Efficiency o 1.5V Start-Up Voltage o 85A Max Total Quiescent Supply Current o 1A Shutdown Mode o Independent Shutdown Inputs o Drives Surface-Mount, Dual N-Channel MOSFETs o Low-Battery Input/Output Comparator o Step-Up/Down Configurable
_______________General Description
The MAX863 dual-output DC-DC converter contains two independent step-up controllers in a single compact package. This monolithic bi-CMOS design draws only 85A when both controllers are on. The input range extends down to 1.5V, permitting use in organizers, translators, and other low-power hand-held products. The MAX863 provides 90% efficiency at output loads from 20mA to over 1A. This space-saving device is supplied in a 16-pin QSOP package that fits in the same area as an 8-pin SOIC. The device uses a current-limited, pulse-frequencymodulated (PFM) control architecture that reduces startup surge currents and maintains low quiescent currents for excellent low-current efficiency. Each controller drives a low-cost, external, N-channel MOSFET switch, whose size can be optimized for any output current or voltage. In larger systems, two MAX863s can be used to generate 5V, 3.3V, 12V, and 28V from just two or three battery cells. An evaluation kit (MAX863EVKIT) is available to speed designs. For a single-output controller, refer to the MAX608 and MAX1771 data sheets.
MAX863
______________Ordering Information
PART MAX863C/D MAX863EEE TEMP. RANGE 0C to +70C -40C to +85C PIN-PACKAGE Dice* 16 QSOP
*Dice are tested at TA = +25C.
________________________Applications
2- and 3-Cell Portable Equipment Organizers Translators Hand-Held Instruments Palmtop Computers Personal Digital Assistants (PDAs) Dual Supply (Logic and LCD)
__________Typical Operating Circuit
VIN
OUT1 SENSE1 VDD BOOT N EXT1 CS1 EXT2 CS2 N
OUT2
__________________Pin Configuration
TOP VIEW
SENSE1 1 VDD 2 FB1 3 BOOT 4 CS1 5 EXT1 6 GND 7 PGND 8 16 REF 15 SHDN2 14 LBI
MAX863
LBO LOW-BATTERY DETECTOR OUTPUT LBI FB2 SHDN1 SHDN2 REF FB1 PGND GND ON/OFF
MAX863
13 LBO 12 FB2 11 SHDN1 10 CS2 9 EXT2
QSOP ________________________________________________________________ Maxim Integrated Products 1
For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
ABSOLUTE MAXIMUM RATINGS
VDD to GND ............................................................-0.3V to +12V PGND to GND .......................................................-0.3V to +0.3V SHDN1, SHDN2, SENSE1, LBO to GND ................-0.3V to +12V EXT1, EXT2 to PGND..................................-0.3V to (VDD + 0.3V) FB1, FB2, CS1, CS2, SEL, LBI, BOOT to GND.................................-0.3V to (VDD + 0.3V) LBO Continuous Output Current.........................................15mA EXT1, EXT2 Continuous Output Current .............................50mA Continuous Power Dissipation (TA = +70C) QSOP (derate 8.30mW/C above +70C) ...................667mW Operating Temperature Range MAX863EEE ....................................................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +160C Lead Temperature (soldering, 10sec) .............................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VDD = +5V, ILOAD = 0mA, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.) PARAMETER VDD Input Voltage OUT1 Output Voltage (Note 3) Quiescent Current Shutdown Current Load Regulation Line Regulation FB1, FB2, LBI Threshold Voltage (Note 4) FB1, FB2, LBI Input Current SHDN1, SHDN2, SEL, BOOT Input High Voltage SHDN1, SHDN2, SEL, BOOT Input Low Voltage SHDN1, SHDN2, SEL, BOOT Input Current CS1, CS2 Threshold Voltage CS1, CS2 Input Current Maximum Switch On-Time Minimum Switch Off-Time EXT Rise/Fall Time (Note 5) EXT On-Resistance LBO Leakage Current LBO Low Level ILBO VLBO,L VLBO = 11V, VLBI > 1.275V ILBO,SINK = 1mA, VLBI < 1.225V 0.1 tON tOFF CLOAD = 1nF, 10% to 90% 14 1.6 VFB, VLBI IFB, ILBI VIH VIL II VCS 2.7V < VDD < 11V VDD = 1.5V 2.7V < VDD < 11V VDD = 1.5V Logic input = VDD or GND 85 100 1 17.5 2 50 5 1 0.4 1.6 0.7 x VDD 0.4 0.2 x VDD 1 115 25 22 2.4 SYMBOL VDD VOUT1 (Note 2) FB1 = VDD FB1 = GND SHDN1 = SHDN2 = VDD, measured from VDD IDD IDD, SHDN SHDN1 = VDD, SHDN2 = GND, measured from VDD SHDN1 = SHDN2 = GND VIN = 3.3V, VOUT1 = 5V, ILOAD = 0mA to 500mA, Figure 2 VIN = 2.7V to 5V, VOUT1 = 5V, ILOAD = 300mA, Figure 2 1.225 40 8 1.25 2 1.275 10 CONDITIONS VDD = OUT1 = BOOT (Note 1) MIN 1.5 2.7 3.2 4.85 3.3 5 50 35 TYP MAX 11 11 3.4 5.15 85 60 1 A A mV/A mV/V V nA V V A mV A s s ns A V UNITS V V
2
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Dual, High-Efficiency, PFM, Step-Up DC-DC Controller
ELECTRICAL CHARACTERISTICS
(VDD = +5V, ILOAD = 0mA, TA = 0C to +85C, unless otherwise noted.) (Note 6) PARAMETER VDD Input Voltage OUT1 Output Voltage (Note 3) Quiescent Current Shutdown Current FB1, FB2 Threshold Voltage CS1, CS2 Threshold Voltage Note 1: Note 2: Note 3: Note 4: Note 5: Note 6: SYMBOL VDD VOUT1 (Note 2) FB1 = VDD FB1 = GND SHDN1 = SHDN2 = VDD, measured from VDD IDD IDD, SHDN VFB VCS SHDN1 = VDD, SHDN2 = GND, measured from VDD SHDN1 = SHDN2 = GND 1.21 85 CONDITIONS VDD = OUT1 (Note 1) MIN 1.6 2.8 3.15 4.8 TYP MAX 11 11 3.45 5.2 85 60 1 1.285 115 A A V mV UNITS V V
MAX863
When bootstrapped, an internal low-voltage oscillator drives the EXT1 pin rail-to-rail for low supply voltages. For non-bootstrapped operation, VDD > 2.7V is required to allow valid operation of all internal circuitry. For adjustable output voltages, see the Set the Output Voltage section. Measured with LBI falling. Typical hysteresis is 15mV. EXT1 and EXT2 swing from VDD to GND. Specifications to -40C are guaranteed by design and not production tested.
__________________________________________Typical Operating Characteristics
(TA = +25C, unless otherwise noted.)
EFFICIENCY vs. OUTPUT CURRENT (VOUT1 = 3.3V, BOOTSTRAPPED)
MAX863 toc01
EFFICIENCY vs. OUTPUT CURRENT (VOUT1 = 5.0V, BOOTSTRAPPED)
MAX863 toc02
EFFICIENCY vs. OUTPUT CURRENT (VOUT1 = 5.0V, NON-BOOTSTRAPPED)
90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 1 10 VOUT1 = 5.0V A: VIN = 2.7V B: VIN = 3.3V C: VIN = 3.6V D: VIN = 4.0V 100 1000 A B D
MAX863 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 1 10 100 1000 OUTPUT CURRENT (mA) VOUT1 = 3.3V A: VIN = 1.5V B: VIN = 2.4V C: VIN = 2.7V B A C
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 1 10 B A C
100
F
C
D
E
VOUT1 = 5.0V A: VIN = 1.5V B: VIN = 2.4V C: VIN = 2.7V D: VIN = 3.3V E: VIN = 3.6V F: VIN = 4.0V 100 1000
OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
_______________________________________________________________________________________
3
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
____________________________Typical Operating Characteristics (continued)
(TA = +25C, unless otherwise noted.)
EFFICIENCY vs. OUTPUT CURRENT (VOUT1 = 12V, NON-BOOTSTRAPPED)
MAX863 toc04
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 1 10 100 A VOUT1 = 5.0V A: VIN = 2.7V B: VIN = 3.3V C: VIN = 3.6V D: VIN = 4.0V E: VIN = 6.0V B C D E
BOOTSTRAPPED-MODE MINIMUM START-UP INPUT VOLTAGE vs. OUTPUT CURRENT
MAX863toc05
3.5 3.0 VOUT1 = 5V 2.5 2.0 1.5 1.0 0.5 VOUT1 = 3.3V
START-UP INPUT VOLTAGE (V) 1000
1 1 10 100 1000 OUTPUT CURRENT (mA)
OUTPUT CURRENT (mA)
VDD CURRENT vs. VDD VOLTAGE
MAX863 toc15
60 50 VDD CURRENT (A) 40 30 20 10 0 0 2 4 6 8 10 BOTH ON
EXT RISE AND FALL TIMES vs. SUPPLY VOLTAGE AND MOSFET CAPACITANCE
C,1 C,2 A: 470pF B: 1.0nF C: 2.2nF 1: RISE 2: FALL
MAX863 toc07
140 120 RISE/FALL TIME (ns) 100 80 60 40 A,2 20 B,2 A,1 B,1
CONVERTER 1 ON CONVERTER 2 ON
12
0 0 2 4 6 8 10 12 SUPPLY VOLTAGE (V) Cond: Single 5V
VDD VOLTAGE (V) Cond: Single +5V
LOAD-TRANSIENT RESPONSE
MAX863 toc08
RESPONSE ENTERING/ EXITING SHUTDOWN (BOOTSTRAPPED)
A
LINE-TRANSIENT RESPONSE
MAX863 toc09 MAX863 toc10
A B C 3.3V
A B
C B 100s/div VOUT1 = 3.3V, IOUT1 = 100mA TO 600mA A: VOUT1, 100mV/div, 3.3V DC OFFSET B: IOUT1, 200mA/div 200s/div VOUT1 = 3.3V, IOUT1 = 100mA, VIN = 2.4V A: SHDN1, 5V/div B: INDUCTOR CURRENT, 2A/div C: VOUT1, 3.3V OFFSET, 500mV/div 500s/div VOUT1 = 5V, IOUT1 = 800mA A: VIN = 2.7V TO 3.7V, 500mV/div B: VOUT1, AC COUPLED, 50mV/div C: INDUCTOR CURRENT, 2A/div
0A
4
_______________________________________________________________________________________
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller
______________________________________________________________Pin Description
PIN 1 2 3 NAME SENSE1 VDD FB1 FUNCTION Feedback Input for DC-DC Controller 1 in Fixed-Output Mode IC Power-Supply Input Adjustable Feedback and Preset Output Voltage Selection Input for DC-DC Controller 1. Connect to VDD for 3.3V preset output or to GND for 5V output. Connect a resistor voltage divider to adjust the output voltage. See the section Set the Output Voltage. Bootstrap Low-Voltage-Oscillator Enable Input. BOOT is an active-high, logic-level input. It enables the low-voltage oscillator to allow start-up from input voltages down to 1.5V while in a bootstrapped circuit configuration. Connect BOOT to GND when in a non-bootstrapped configuration. If BOOT is high, VDD must be connected to OUT1. Input to the Current-Sense Comparator of DC-DC Controller 1 Gate-Drive Output of DC-DC Controller 1. Drives an external N-channel power MOSFET. Analog Ground for Internal Reference, Feedback, and Control Circuits High-Current Ground Return for Internal MOSFET Drivers Gate-Drive Output of DC-DC Controller 2. Drives an external N-channel power MOSFET. Input to the Current-Sense Amplifier of DC-DC Controller 2 Active-Low Shutdown Input for DC-DC Controller 1. Connect to VDD for normal operation. Adjustable Feedback Input for DC-DC Controller 2. Connect a resistor voltage divider to adjust the output voltage. See the section Set the Output Voltage. Low-Battery Output. An open-drain N-channel MOSFET output. Sinks current when the voltage on LBI drops below 1.25V. If unused, connect to GND. Low-Battery Comparator Input. When the voltage on LBI drops below 1.25V, LBO sinks current. If unused, connect to GND. Active-Low Shutdown Input for DC-DC Controller 2. Connect to VDD for normal operation. Reference Bypass Input. Connect a 0.1F ceramic capacitor from REF to GND.
MAX863
4
BOOT
5 6 7 8 9 10 11 12 13 14 15 16
CS1 EXT1 GND PGND EXT2 CS2 SHDN1 FB2 LBO LBI SHDN2 REF
_______________Detailed Description
The MAX863 dual, bi-CMOS, step-up, switch-mode power-supply controller provides preset 3.3V, 5V, or adjustable outputs. Its pulse-frequency-modulated (PFM) control scheme combines the advantages of low supply current at light loads and high efficiency with heavy loads. These attributes make the MAX863 ideal for use in portable battery-powered systems where small size and low cost are extremely important, and where low quiescent current and high efficiency are needed to maximize operational battery life. Use of external current-sense resistors and MOSFETs allows the designer to tailor the output current and voltage capability for a diverse range of applications.
PFM Control Scheme
Each DC-DC controller in the MAX863 uses a one-shotsequenced, current-limited PFM design, as shown in Figure 1. Referring to the Typical Operating Circuit (Figure 2) and the switching waveforms (Figures 3a-3f), the circuit works as follows. Output voltage is sensed by the error comparator using either an internal voltage divider connected to SENSE1 or an external voltage divider connected to FB1. When the output voltage drops, the error comparator sets an internal flip-flop. The flip-flop turns on an external MOSFET, which allows inductor current to ramp-up, storing energy in a magnetic field.
_______________________________________________________________________________________
5
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Q REF TRIG
FB2 ERROR COMPARATOR
MAX ON-TIME ONE-SHOT
MAX863
S Q TRIG MIN ON-TIME ONE-SHOT Q R TIMING BLOCK EXT2
CS2
CURRENT-SENSE COMPARATOR
100mV CS1 CURRENTSENSE COMPARATOR
PGND
SENSE1
VDD - 100mV
LOWVOLTAGE OSCILLATOR
BOOT
EXT1 100mV FB1 UVLO VDD TIMING BLOCK
100mV LBO REF N LBI N
REF
ERROR COMPARATOR VDD
REF 1.25V BIAS
GND
REF
SHDN2
SHDN1
Figure 1. Functional Diagram
The flip-flop resets and turns off the MOSFET when either a) the voltage across the current-sense resistor exceeds 100mV, or b) the 17.5s maximum on-time one-shot trips. When the MOSFET turns off, the magnetic field begins to collapse, and forces current into the output capacitor and load. As the stored energy is transferred to the output, the inductor current ramps down. The output capacitor smoothes out the energy transfer by storing charge when the diode current is
6
high, then supplying current to the load during the first half of each cycle, maintaining a steady output voltage. Resetting the flip-flop sets the off-time one-shot, disabling the error-comparator output and forcing the MOSFET off for at least 2s to enforce a minimum time for energy transfer to the output. The MAX863 waits until the output voltage drops again before beginning another cycle. The MAX863's switching frequency depends on the load current and input voltage.
_______________________________________________________________________________________
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
VIN = 1.5V TO THE LOWER OF VOUT1 OR VOUT2 C3 100F 10V 0.1 C4 100F 10V 0.1
L1 10H 2A VOUT1 = 5V C1 220F 10V 0.1 R5 C2 0.1F D1 MBRS340T3
L2 10H 2A D2 MBRS340T3 VOUT2 = 3.3V C5 330F 10V 0.1 R3 165k 1% C6 10pF
SENSE1 N1A EXT1 CS1
VDD
BOOT EXT2 CS2 R2 50m N1B IRF7301
R7 100k
R1 50m
MAX863
LBO FB2 SHDN1 LBI SHDN2 REF FB1 PGND GND C7 0.1F ON/OFF
LOW-BATTERY DETECTOR OUTPUT
R4 100k 1%
R6
Figure 2. Bootstrapped Typical Operating Circuit
Continuous/Discontinuous-Conduction Modes
Each converter in the MAX863 determines from moment to moment whether to switch or not, waiting until the output voltage drops before initiating another cycle. Under light loads, the inductor current ramps to zero before the next cycle; this is discontinuous-conduction mode. Continuous-conduction mode occurs when the next switching cycle begins while current is still flowing through the inductor. The transition point between discontinuous- and continuous-conduction mode is determined by input and output voltages, and by the size of the inductor relative to the peak switching current. In general, reducing inductance toward the minimum recommended value pushes the transition point closer to the maximum load current. If the inductor value is low enough or the output/input voltage ratio high enough, the DC-DC converter may remain in discontinuous-conduction mode throughout its entire load range.
The MAX863 transitions into continuous-conduction mode in two ways, depending on whether preset or adjustable mode is used and how the external feedback network is compensated. Under light loads, the IC switches in single pulses (Figure 3a). The threshold of transition into continuous-conduction mode is reached when the inductor current waveforms are adjacent to one another, as shown in Figure 3b. As the load increases, the transition into continuous-conduction mode progresses by raising the minimum inductor current (Figures 3c, 3d). Depending on feedback compensation, transition into continuous-conduction mode may also progress with grouped pulses (Figures 3e, 3f). Pulse groups should be separated by less than two or three switching cycles. Output ripple should not be significantly more than the single-cycle no-load case.
_______________________________________________________________________________________
7
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
A OV
B
3.3V
C 20s/div a) IOUT1 = 287mA 20s/div b) IOUT1 = 608mA 20s/div c) IOUT1 = 767mA 20s/div d) IOUT1 = 1.01A
0A
A VOUT1 = 3.3V PLOTS a-d: INTERNAL FEEDBACK PLOTS e-f: UNCOMPENSATED, EXTERNAL FEEDBACK A: MOSFET DRAIN, 2V/div B: VOUT1, 100mV/div, 3.3V DC OFFSET C: INDUCTOR CURRENT, 1A/div
OV
B
3.3V
C 20s/div e) IOUT1 = 757mA 20s/div f) IOUT1 = 881mA
0A
Figures 3a-3f. MAX863 Switching Waveforms During Transition into Continuous Conduction
VIN = 2.7V TO THE LOWER OF VOUT1 OR VOUT2 C3 100F L1 10V 10H 0.1 2A SENSE1 N1.A EXT1 CS1 R5 R7 100k R1 50m C4 C2 100F 0.1F 10V 0.1
D1 MBRS340T3 VOUT1 = 5V C1 220F 10V 0.1
L2 10H 2A D2 MBRS340T3
VOUT2 = 12V C5 100F 20V 0.1
VDD EXT2 CS2 1M R2 50m R3 1M 1% C6 10pF N1.B IRF7301
MAX863
LBO FB2 SHDN1 LBI SHDN2 REF FB1 BOOT PGND GND C7 0.1F ON/OFF
LOW-BATTERY DETECTOR OUTPUT
R6
R4 115k 1%
C8 82pF
Figure 4a. Non-Bootstrapped Typical Operating Circuit
8 _______________________________________________________________________________________
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
VIN = 2.7V TO 11V C3 100F 20V 0.1 C2 0.1F C4 100F 20V 0.1
L1 10H 2A D1 MBRS340T3 R5
L2 10H 1A D2 MBRS140
R6
VOUT1 = 12V C1 100F 16V 0.1 C8 10pF R8 1M 1%
VOUT2 = 24V C5 22F 35V 0.1
VDD N1.A EXT1 CS1 R7 100k R1 50m
LBI EXT2 CS2 R2 100m R3 1M 1% C6 15pF N1.B IRF7301
MAX863
LBO LOW-BATTERY DETECTOR OUTPUT FB1 FB2 SHDN1 SHDN2 REF PGND GND C7 0.1F ON/OFF
C9 82pF
R9 115k 1%
SENSE1 BOOT
R4 56k 1%
C10 270pF
Figure 4b. Adjustable Non-Bootstrapped Typical Operating Circuit
Low-Voltage Start-Up Oscillator (BOOT Pin)
The MAX863 features a low-voltage start-up oscillator that guarantees start-up in bootstrapped configuration down to 1.5V. At these low supply voltages, the error comparator and internal biasing of the IC are locked out. The low-voltage oscillator switches the external MOSFET with around 30% duty cycle until the voltage at VDD rises above 2.7V. At this point, the error comparator and one-shot timing circuitry turn on. The lowvoltage oscillator is enabled by connecting the BOOT pin to VDD. When BOOT is high, VDD must be connected to VOUT1. Use the start-up oscillator in the bootstrapped configuration only, since the MAX863 operates in an open-loop state while the start-up oscillator is active. When using a non-bootstrapped circuit configuration, connect BOOT to GND to disable the start-up oscillator. This prevents the output from rising too high when VDD is between 1.5V and 2.7V, such as during power-up and low-battery conditions.
Bootstrapped/Non-Bootstrapped Modes
Figures 2 and 4 show standard applications in bootstrapped and non-bootstrapped modes. In boot-
strapped mode, the IC is powered from the output (VDD is connected to OUT1, BOOT is connected to VDD). Bootstrapped-mode operation is useful for increasing the gate drive to the MOSFETs in low-input-voltage applications, since EXT1 and EXT2 swing from VDD to GND. Increasing the gate-drive voltage reduces MOSFET on-resistance, which improves efficiency and increases the load range. For supply voltages below 5V, bootstrapped mode is recommended. In bootstrapped mode, the output connected to VDD must not exceed 11V. If BOOT is high, VDD must be connected to OUT1. In non-bootstrapped mode, the IC is powered by a direct connection from the input voltage to VDD. Since the voltage swing applied to the gate of the external MOSFET is derived from VDD, the external MOSFET onresistance increases at low input voltages. The minimum input voltage is 2.7V. For operation down to 4V, use logic-level MOSFETs. For lower input voltages, use low-threshold logic-level MOSFETs. When both output voltages are set above 11V, non-bootstrapped mode is mandatory.
_______________________________________________________________________________________
9
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Shutdown Mode
The MAX863 has two shutdown inputs useful for conserving power and extending battery life. Driving SHDN1 or SHDN2 low turns off the corresponding DCDC controller and reduces quiescent current. Driving both shutdown pins low turns off the reference, control, and biasing circuitry, putting the MAX863 in a 1A shutdown mode. Connect SHDN1 and SHDN2 to VDD for normal operation. tested prior to production. Table 1 provides a list of component suppliers. Two design methods are included. The first uses graphs for selecting the peak current required for 3.3V, 5V, 12V, and 24V outputs. The second uses equations for selecting the peak current and inductor value in circuits with other outputs. When designing high-voltage, flyback, SEPIC, and autotransformer boost circuits, contact Maxim's Applications Department for the appropriate design equations.
__________________Design Procedure
Boost DC-DC converters using the MAX863 can be designed in a few simple steps to yield a working firstiteration design. All designs should be prototyped and
Specify Design Objectives
For each of the two outputs, specify the output voltage and maximum load current, as well as maximum and
MAX863 FIG05A
MAXIMUM OUTPUT CURRENT (A)
2.0
MAXIMUM OUTPUT CURRENT (A)
1.5
VOUT = 3.3V, L = 1.5 LMIN A: IPEAK = 3A B: IPEAK = 2A C: IPEAK = 1.47A D: IPEAK = 1A E: IPEAK = 0.67A F: IPEAK = 0.5A
A
1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 1.0
B C D
VOUT = 5V, L = 1.5 LMIN A: IPEAK = 3A B: IPEAK = 2A C: IPEAK = 1.47A D: IPEAK = 1A E: IPEAK = 0.67A F: IPEAK = 0.5A
A
B C D E F
1.0
0.5
E F
0 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 INPUT VOLTAGE (V) Cond: Single +5V Code = FFFhex
1.5
2.0
2.5
3.0
3.5
4.0
4.5
INPUT VOLTAGE (V) Cond: Single +5V Code = FFFhex
Figure 5a. Maximum Output Current vs. Input Voltage and IPEAK (VOUT = 3.3V)
Figure 5b. Maximum Output Current vs. Input Voltage and IPEAK (VOUT = 5V)
MAX863 FIG05C
MAXIMUM OUTPUT CURRENT (A)
2.0 1.5
MAXIMUM OUTPUT CURRENT (A)
VOUT = 12V, L = 1.5 LMIN A: IPEAK = 3A B: IPEAK = 2A C: IPEAK = 1.47A D: IPEAK = 1A E: IPEAK = 0.67A F: IPEAK = 0.5A
A
0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0
B C D
VOUT = 24V, L = 1.5 LMIN A: IPEAK = 3A B: IPEAK = 2A C: IPEAK = 1.47A D: IPEAK = 1A E: IPEAK = 0.67A F: IPEAK = 0.5A
A
B C D
1.0
0.5
E F
E F 0 2 4 6 8 10 12
0 0 2 4 6 8 10 12 INPUT VOLTAGE (V) Cond: Single +5V Code = FFFhex
INPUT VOLTAGE (V) Cond: Single +5V Code = FFFhex
Figure 5c. Maximum Output Current vs. Input Voltage and IPEAK (VOUT = 12V)
10
Figure 5d. Maximum Output Current vs. Input Voltage and IPEAK (VOUT = 24V)
______________________________________________________________________________________
MAX863 FIG05D
2.5
1.0
MAX863 FIG05B
2.5
2.0
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller
minimum input voltages. Estimate the maximum input currents for each output based on the minimum input voltage and desired output power: 0.8 x VIN(MIN)
INDUCTOR CURRENT, IL
MAX863
IIN,DC(MAX)
VOUT x IOUT
MIN =
IL IPEAK
where 0.8 is chosen as a working value for the nominal efficiency. The power source must be capable of delivering the sum of the maximum input currents of both DC-DC converters.
IL IPEAK
Determine the Peak Switching Current (Graphical Method)
The peak switching current set by RSENSE determines the amount of energy transferred from the input on each cycle. For 3.3V, 5V, 12V, and 24V output circuits, the peak current can be selected using the output current curves shown in Figures 5a-5d.
t
Figure 6. Ratio of Inductor AC Ripple Current to Peak Current
Determine the Peak Switching Current and Inductance (Analytical Method)
The following boost-circuit equations are useful when the desired output voltage differs from those listed in Figure 5. They allow trading off peak current and inductor value in consideration of component availability, size, and cost. Begin by calculating the minimum allowable ratio of inductor AC ripple current to peak current, MIN (Figure 6): MIN = t OFF(MIN) t ON(MAX ) x VOUT - VIN(MIN) VIN(MIN)
The suggested inductor value is: L V OUT - VIN(MIN) x tOFF(MIN) IPEAK x
Round L up to the next standard inductor value.
Choose RSENSE The peak switching current is set by RSENSE (R1 and R2 in Figure 2):
RSENSE VCS(MIN) IPEAK = 85mV IPEAK
where tOFF(MIN) = 2s and tON(MAX) = 17.5s. Select a value for greater than MIN. If MIN is less than 1, an acceptable choice is (MIN + 1) / 2. If MIN is greater than 1, values between MIN and 2 x MIN are acceptable (1.5 x MIN, for example). Values greater than 1 represent designs with full-load operation in discontinuous-conduction mode. Now calculate the peak switching current and inductance. If MIN 1, use: IPEAK = IIN,DC(MAX ) x For 1%, use: VOUT + VIN x - 1 IPEAK = 2 x IIN,DC(MAX ) x VOUT 2 2-
Verify that you've selected the correct RSENSE by testing the prototype using the minimum input voltage while supplying the maximum output current. If the output voltage droops, then decrease the value of the current-sense resistor and adjust the other components as necessary. The current-sense resistor must be a small, low-inductance type such as a surface-mount metal-strip resistor. Do not use wire-wound resistors, since their high inductance will corrupt the current feedback signal. In order to allow use of standard resistor values, round RSENSE to the next lowest value. The current-sense resistor's power rating should be higher than: RPOWER RATING = V CS(MAX) RSENSE
2
(
)
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11
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Select the Inductor Component
Two essential parameters are required for selecting the inductor: inductance and current rating. Inductance should be low enough to allow the MAX863 to reach the peak current limit during each cycle before the 17.5s maximum on-time. Conversely, if the inductance is too low, the current will ramp up to a high level before the current-sense comparator can turn the switch off. A practical minimum on-time (tON(MIN)) is 1.5s. LMIN and: LMAX VIN(MIN) x t ON(MAX) IPEAK VIN(MAX) x t ON(MIN) IPEAK MOSFETs when starting from input voltages below 4V. This also applies in bootstrapped mode to ensure start-up. The MOSFET in a simple boost converter must withstand the output voltage plus the diode forward voltage. Voltage ratings in SEPIC, flyback, and autotransformer-boost circuits are more stringent. Choose a MOSFET with a maximum continuous draincurrent rating higher than the current limit set by CS. The two most significant losses contributing to the MOSFET's power dissipation are I2R losses and switching losses. Reduce I2R losses by choosing a MOSFET with low RDS(ON), preferably near the current-sense resistor value or lower. A MOSFET with a gate charge (Qg) of 50nC or smaller is recommended for rise and fall times less than 100ns on the EXT pins. Exceeding this limit results in slower MOSFET switching speeds and higher switching losses, due to a longer transition time through the linear region as the MOSFET turns on and off.
When selecting IPEAK using the graphs in Figure 5, choose inductance values between 1.3 and 1.7 times the minimum inductance value to provide a good tradeoff between switching frequency and efficiency. The lower of the inductor saturation current rating or heating current rating should be greater than IPEAK: ISATURATION and IHEATING > IPEAK The saturation current limit is the current level where the magnetic field in the inductor has reached the maximum the core can sustain, and inductance starts to fall. The heating current rating is the maximum DC current the inductor can sustain without overheating. Disregarding the inductor's saturation current rating is a common mistake that results in poor efficiency, bad regulation, component overheating, or other problems. The resistance of the inductor windings should be comparable to or less than that of the current-sense resistor. To minimize radiated noise in sensitive applications, use a toroid, pot core, or shielded bobbin core inductor.
Select the Output Diode
Schottky diodes, such as the 1N5817-1N5822 family or surface-mount equivalents, are recommended. Ultrafast silicon rectifiers with reverse recovery times around 60ns or faster, such as the MUR series, are acceptable but have greater forward voltage drop. Make sure that the diode's peak current rating exceeds the current limit set by RSENSE, and that its breakdown voltage exceeds VOUT. Schottky diodes are preferred for heavy loads, especially in low-voltage applications, due to their low forward voltage. For high-temperature applications, some Schottky diodes may be inadequate due to high leakage currents. In such cases, ultra-fast silicon rectifiers are recommended, although acceptable performance can often be achieved by using a Schottky diode with a higher reverse voltage rating.
Determine Input and Output Filter Capacitors
Low-ESR capacitors are recommended for both input bypassing and output filtering. Capacitor equivalent series resistance (ESR) is a major contributor to output ripple--typically 60% to 90%. Low-ESR tantalum capacitors offer a good tradeoff between price and performance. Ceramic and Sanyo OS-CON capacitors have the lowest ESR. Ceramic capacitors are often a good choice in high-output-voltage applications where large capacitor values may not be needed. Low-ESR aluminum-electrolytic capacitors are tolerable and can be used when cost is the primary consideration; however, standard aluminum-electrolytic capacitors should be avoided.
Choose the MOSFET Power Transistor
Use N-channel MOSFETs with the MAX863. When selecting an N-channel MOSFET, five important parameters are gate-drive voltage, drain-to-source breakdown voltage, current rating, on-resistance (RDS(ON)), and total gate charge (Qg). The MAX863's EXT1 and EXT2 outputs swing from GND to VDD. To ensure the external N-channel MOSFET is turned on sufficiently, use logic-level MOSFETs when VDD is less than 8V and low-threshold logic-level
12
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Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Table 1. Component Suppliers
VOUT1 OR VOUT2
SUPPLIER Inductors Coilcraft
PHONE (847) 639-6400 (561) 241-7876 (605) 668-4131 (847) 956-0666 (516) 435-1110 (310) 322-3331 (602) 303-5454 (402) 564-3131 (512) 992-7900 (803) 946-0690 (619) 661-6835 (603) 224-1961
FAX (847) 639-1469 (561) 241-9339 (605) 665-1627 (847) 956-0702 (516) 435-1824 (310) 322-3232 (602) 994-6430 (402) 563-6418 (512) 992-3377 (803) 626-3123 (619) 661-1055 (603) 224-1430
R1
C1 (OPTIONAL)
COUT
Coiltronics Dale Inductors Sumida USA MOSFETs and Diodes Central Semiconductor International Rectifier
MAX863
FB1 OR FB2
R2
C2 (OPTIONAL FOR HIGHVOLTAGE CIRCUITS)
Motorola Current-Sense Resistors Dale/Vishay IRC Electrolytic Capacitors AVX Sanyo USA Sprague
Figure 7. Adjustable Output Circuit
Large-Value Ceramic Capacitors Marcon/United Chemi-Con TDK Vishay/Vitramon (847) 696-2000 (847) 390-4373 (203) 268-6261 (847) 696-9278 (847) 390-4428 (203) 452-5670
Voltage ripple is the sum of contributions associated with ESR and the capacitor value, as shown below: VRIPPLE VRIPPLE,ESR + VRIPPLE,C To simplify selection, assume that 75% of the ripple results from ESR and that 25% results from the capacitor value. Voltage ripple as a consequence of ESR is approximated by: VRIPPLE,ESR RESR x IPEAK so:
Add VDD and REF Bypass Capacitors
Bypass the MAX863 with 0.1F or higher value ceramic capacitors placed as close to the VDD, REF, and GND pins as possible.
Set the Output Voltage
RESR VRIPPLE,ESR IPEAK
DC-DC converter 1 operates with a 3.3V, 5V, or adjustable output. For a preset output, connect SENSE1 to OUT1 (Figures 2 and 4a), then set FB1 to VDD for 3.3V operation or to GND for 5V operation. For an adjustable output, connect a resistor voltage divider to the FB1 pin (Figure 7). In adjustable output circuits, connect SENSE1 to GND. DC-DC converter 2 can be adjusted from very high voltages down to VIN using external resistors connected to the FB2 pin, as shown in Figure 7. Select feedback resistor R2 in the 10k to 500k range. R1 is given by: V R1 = R2 OUT - 1 1.25V where 1.25V is the voltage of the internal reference.
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Estimate input and output capacitor values for a given voltage ripple as follows:
C
0.5L x I2PEAK VRIPPLE,C x V
where V is the input or output voltage, depending on which capacitor is being calculated. Choose input capacitors with working voltage ratings over the maximum input voltage, and output capacitors with working voltage ratings higher than their respective outputs.
Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Set Feedback Compensation
External voltage feedback to the MAX863 should be compensated for stray capacitance and EMI in the feedback network. Proper compensation is achieved when the MAX863 switches evenly, rather than in widely spaced bursts of pulses with large output ripple. Typically, lead compensation consisting of a 10pF to 220pF ceramic capacitor (C1 in Figure 7) across the upper feedback resistor is adequate. Circuits with VOUT or VDD greater than 7.5V may require a second capacitor across the lower feedback resistor. Initially, choose this capacitor so that R2C2 = R1C1. Set the final values of the compensation capacitors based on empirical analysis of a prototype. corrupting the voltage and current-feedback signals. Place power components as close together as possible, and keep their traces short, direct, and wide. Keep the extra copper on the board and integrate it into ground as an additional plane. On multi-layer boards, avoid interconnecting the ground pins of the power components using vias through an internal ground plane. Instead, place the ground pins of the power components close together and route them in a "star" ground configuration using component-side copper, then connect the star ground to the internal ground plane using multiple vias. The current-sense resistor and voltage-feedback networks should be very close to the MAX863. Noisy traces, such as from the EXT pins, should be kept away from the voltage-feedback networks and isolated from them using grounded copper. Consult the MAX863 evaluation kit manual for a full PC board example.
PC Board Layout and Routing
High switching speeds and large peak currents make PC board layout an important part of design. Poor layout can cause excessive EMI and ground-bounce, both of which can cause instability or regulation errors by
VIN = 1.8V TO VOUT1 C3 100F 10V 0.1 C4 100F 10V 0.1
D1 MBRS340T3 VOUT1 = 5V C1 220F 10V 0.1 R5
L1 10H 2A
L2 10H 1A D2 MBRS140 VOUT2 = 24V, 35mA C5 22F 35V 0.1 R3 909k 1% C6 15pF
SENSE1 C2 0.1F N1A EXT1 CS1 R7 100k R1 50m
VDD
BOOT EXT2 CS2 R2 100m N1B IRF7103
MAX863
LBO FB2 SHDN1 LBI SHDN2 REF FB1 PGND GND C7 0.1F ON/OFF
LOW-BATTERY DETECTOR OUTPUT
R6
R4 49.9k 1%
C8 270pF
Figure 8. Bootstrapped 3.3V Logic and 24V LCD Bias Supply
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Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
__________Applications Information
Low-Input-Voltage Operation
When the voltage at V DD falls and EXT1 or EXT2 approaches the MOSFET gate-to-source threshold voltage, the MOSFET may operate in its linear region and dissipate excessive power. Prolonged operation in this mode may damage the MOSFET if power dissipation ratings are inadequate. This effect is more significant in non-bootstrapped mode, but can occur in bootstrapped mode if the input voltage drops so low that it cannot support the load and causes the output voltage to collapse. To avoid this condition, use logic-level or low-threshold MOSFETs.
Starting Up Under Load
The Typical Operating Characteristics show the Bootstrapped-Mode Minimum Start-Up Input Voltage vs. Output Current graph. The MAX863 is not intended to start up under full load in bootstrapped mode with low input voltages.
________________Application Circuits
Bootstrapped 5V Logic and 24V LCD Bias Supply
The circuit in Figure 8 operates from two AA or AAA cells, and generates 5V (up to 750mA) for logic and 24V (up to 35mA) for an LCD bias supply. OUT1 is used to bootstrap the MAX863 for better MOSFET gate drive. V OUT1 can be set to 3.3V if low threshold MOSFETs are used.
VIN = 2.0V TO 11V OR VOUT2 T1 10H, 2.5A CTX10-4 FLYBACK OR SEPIC OUTPUT VOUT1 = 3.3V, 600mA C9 D1 MBRS340T3 10F C3 100F 10V 0.1 C4 100F 10V 0.1 C2 1F D2 MBRS340T3 SENSE1 FB1 N1A EXT1 CS1 R1 50m R5 VDD BOOT EXT2 CS2 R2 50m R3 1M 1% C6 10pF N1B IRF7301 VOUT2 = 12V C5 100F 20V 0.1 L2 10H 2A
D3 CMPSH-3C
C1 330F 10V 0.1
MAX863
R7 100k LBI FB2 R6 SHDN1 SHDN2 LBO LOW-BATTERY DETECTOR OUTPUT PGND GND REF C7 0.1F ON/OFF
R4 115k 1%
C8 82pF
Figure 9. 3-Cell to 3.3V Step-Up/Step-Down Logic Supply with 12V for Flash Memory or Analog Functions
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Dual, High-Efficiency, PFM, Step-Up DC-DC Controller MAX863
Step-Up/Down SEPIC Converter and 12V Supply
The circuit in Figure 9 provides a buck/boost function for applications where the input voltage range can be greater than or less than VOUT1. It provides 3.3V (up to 600mA) or 5V, as well as 12V (up to 200mA at VIN = 2.4V) for powering flash memory or analog functions. The main output employs a SEPIC topology using a coupled inductor and a capacitor to transfer energy to the output. C2 must be a low-ESR type capable of withstanding high ripple current. Ceramic and Sanyo OS-CONs work well, but low-ESR aluminum electrolytics (which are less costly) are tolerable. Do not use a tantalum capacitor for C2. C2's voltage rating must be higher than the maximum input voltage. The MOSFET must withstand a voltage equal to the sum of the input and output voltages; i.e., when converting 11V to 3.3V, the MOSFET must withstand 14.3V. The dual Schottky diode D3 bootstraps power to the MAX863, allowing use of the low-voltage start-up oscillator, as well as improved gate-drive voltages during normal operation.
___________________Chip Information
TRANSISTOR COUNT: 858 SUBSTRATE CONNECTED TO GND
________________________________________________________Package Information
QSOP.EPS
16
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